Miniaturised ultra-wideband circular polarised koch fractal crossed dipole array

Abstract

Ultra-wideband (UWB) systems utilise a broad frequency spectrum to achieve high data transmission rates and enhanced precision, creating a demand for specialised UWB antennas. Circularly Polarised (CP) antennas are considered essential for maintaining robust performance under varied environmental conditions. Among them, the crossed dipole is effective for UWB CP applications, but its relatively large size could limit its use in various systems. A miniaturised crossed dipole design employing the Koch fractal structure is proposed and the measurement results are presented. The prototype antenna achieved an effective bandwidth covering 88% of the frequency range from 1.4 to 3.6 GHz. Additionally, a 4 × 4 array using this design was tested in the same frequency band, demonstrating beam-steering capabilities up to 30°. The proposed antenna has demonstrated effective deployment in diverse communication systems and phased array applications within ultra-wideband (UWB) CP environments.

Abbreviations

AEP

active element pattern

AR

axial ratio

ARBW

axial ratio bandwidth.

CP

circularly polarisation

CPA

circularly polarised antennas

DRA

dielectric resonator antennas

LHCP

left hand circular polarisation

UWB

ultra wideband

1 INTRODUCTION

Recently, the demand for Ultra-Wideband (UWB) antennas has increased across various domains, such as wireless and satellite communication systems, attributable to their stable communication properties. [1-4] Consequently, various researches into UWB circularly polarised antennas (CPAs) have been conducted [5-7]. The use of Circularly Polarised (CP) can not only reduce polarisation mismatches caused by natural phenomena such as the Faraday rotation effect but also effectively minimise losses due to multipath fading [8].

To implement a CPA, it is essential to generate polarisations of equal magnitude with a 90° phase difference. CPAs can be categorised into designs utilising a single feed and those using multiple feeds. By employing more than one feed, it is possible to realising a UWB CP single antenna using sequential rotation feed network that introduces a 90° phase shift to four radiators exhibiting linear polarisation characteristics. However, this approach tends to increase the overall size of the antenna and complicates the feed structure [9, 10]. Designing the antenna with a single feed allows for a relatively simpler structure. A typical example is the patch type antenna. Patch antennas are characterised by simple design and manufacturing processes, compact size, and low-profile features. Generally, to create CP using a patch, various patch shapes are utilised, and a wideband CP is designed by forming slots in the ground. However, UWB CPAs designed with patches tend to have reduced gain due to the back radiation and the asymmetrical shape can distort the radiation pattern [11, 12]. In the case of DRA (Dielectric Resonator Antennas), they can be designed with a simple structure while also providing high radiation efficiency. However, they generally have a relatively narrow operating band. The Stair-Shaped Dielectric Resonator Antenna [13] has a bandwidth of 22%, and the Trapezoidal Dielectric Resonator Antenna [14] offers a bandwidth of 21.5%. Crossed dipoles feature a wide band and can be designed with a simple single-feed structure. Specifically, to implement a wide band, the shape of the dipole forming the crossed dipole part can be variably extended, such as in bowtie or elliptical forms [15, 16]. Additionally, to secure high gain characteristics and a wide band, a cavity structure can be formed behind the crossed dipole [17-19]. Methods that secure a wide band by forming side metallic walls or pillars have also been introduced [20-22]. However, crossed dipoles with a cavity structure inherently have a large size. As the size of the antenna is a critical factor during the design process of wireless communication and other systems further miniaturisation of the antenna is necessary for application in various systems. There is a significant demand for antennas that are smaller yet have the same specifications due to spatial constraints in the design of wireless communication systems [23, 24]. When forming arrays, the size constraints make it difficult to use large antennas as elements. Particularly in designing UWB phased arrays for beam steering, it is necessary to reduce the size of individual antenna elements to avoid generating grating lobes. Utilising Tightly Coupled Arrays can effectively form Ultra-Wideband (UWB) phased arrays; however, the complex structure leads to intricate design and fabrication processes [25]. A crossed dipole can achieve broadband characteristics with a relatively simple structure. To form a phased array capable of beam steering, it is necessary to reduce its size.

Various methods can be considered for the miniaturisation of antennas, among which the fractal structure is a prominent method. Applying a fractal structure can expand the useable band or shift the operating band to lower frequencies [26, 27]. This paper applies the Koch fractal structure to a crossed dipole to miniaturise the single element. Subsequently, an extended metallic side wall was added to obtain a stable axial ratio (AR) over a wide band, and slots were formed at the junction between the sidewall and the ground. The designed single element antenna was prototyped and measured for validation. An array utilising the miniaturised single element with the sequential rotation method is followed. The technique of sequential rotation was implemented through sequential divider, and the array's performance was validated in a 4 × 4 setup. The performance of the designed array was verified through measurements of active S-parameters and active element patterns, confirming its beam-steering capability of up to 30°. The following contents describes the simulation and measurement results for the designed single antenna and array.

2 SINGLE ELEMENT DESIGN

The crossed dipole was designed using a Koch fractal structure. The following sections cover its configuration, parameter study and analysis, and the resulting performance.

2.1 Configuration

The shape of the UWB CPA proposed is shown in Figure 1. The crossed dipole, which is rotated and applied with a Koch fractal structure, is formed on both the top and bottom of the substrate. The feature below the substrate is indicated in yellow. Each crossed dipole is symmetrical about the centre and has been rotated anticlockwise by 15° to secure the AR. The overall size of the proposed antenna was set at 51 mm, and the RF-60A substrate was used, which has a dielectric constant of 6.15 and a loss tangent of 0.0028, to achieve a compact design. The thickness is 1.52 mm. The detailed parameter is shown in Table 1. For the crossed dipole, a circular path was used to generate the necessary phase difference for CP operation. The circular path designed to satisfy a quarter-wavelength of the guide wavelength and has a thickness of 0.5 mm. The crossed dipole features a Koch folded fractal structure. It consists of a 1 mm folded line and has dipole arms of size 21 mm × 4 mm with a single iteration of the Koch fractal. Each fractal is symmetrically centred on the folded line and formed with a width of 4 mm and a height of 1 mm. Fundamentally, the crossed dipole is fed by a 50 Ω coaxial cable; the outer conductor is connected to the dipole below the substrate, and only the inner conductor is connected to the dipole above the substrate. Additionally, to improve the performance of the crossed dipole, a ground reflector spaced 22 mm below the substrate was installed, and a slotted sidewall was positioned above it. The side wall has thickness of 2 mm and slots that are 10 mm high and 4 mm wide to ensure a wider AR bandwidth. For the assembly of the substrate and the cavity in subsequent prototype production, 5 × 5 × 22 mm pillars were added to each corner of the side walls to accommodate screw attachments. The proposed antenna operates as LHCP(Left Hand Circular Polarisation). Due to the element's point-symmetric structure, RHCP can be achieved by making a symmetric modification to the design.

FIGURE 1

Final antenna design configuration: (a) Perspective view. (b) Top view.

TABLE 1. Final antenna design parameter (unit: mm).

Parameter	Value	Parameter	Value	Parameter	Value
g	51	l	26	h	22
sl	4	sh	6	fw	6
fl	21	cw	4	cl	1
kl	19	kw	4	ro	3.52
pl	2	pw	5	ri	4.02

2.2 Parameter study and analysis

To verify the functionality of the Koch folded antenna, results were compared among a simple rectangular patch antenna, a folded dipole without the Koch fractal structure, and a Koch folded dipole, with all processes having the same parameters except for the crossed dipole. Each antenna design is represented in Figure 2, and their performance differences in terms of S₁₁ and AR are detailed in Figure 3. The simple patch-shaped crossed dipole maintains an S₁₁ value below −10 dB up to the 4.3 GHz band and achieves an AR below 3 dB between 4.2 and 4.5 GHz. However, it shows a mismatched AR value, exceeding 3 dB, within the impedance matching bandwidth. Incorporating a folded structure reduces the upper limit of the impedance matching band to 3.8 GHz, while generally improving the AR values throughout the band. Introducing a Koch folded structure results in an AR below 3 dB across the entire operational bandwidth, which extends from 1.5 to 3.7 GHz. Although further increasing the iteration of the Koch fractal structure shifts the operational frequency lower, the impact is minimal; a Koch folded dipole was designed with a single iteration. As the size of the Koch fractal structure is increased, the operating frequency shifts to a lower band. However, the operation in the low-frequency band is fundamentally influenced by the sidewall structure of the cavity, as noted in ref. [22]. Thus, rather than enlarging the Koch fractal structure, its shape was optimised for the operational impedance matching bandwidth bands, resulting in final dimensions of 1 mm in height and 4 mm in width.

FIGURE 2

Illustration of crossed dipole shape change for Koch fractal: (a) Normal patch dipole. (b) Folded dipole. (c) Koch folded dipole.

FIGURE 3

The results of crossed dipole shape change for Koch fractal: (a) S₁₁ parameter. (b) Axial ratio.

To further expand the bandwidth, slots were created in the side walls of the cavity, and the crossed dipole was rotated. The results of the S₁₁ and AR changes due to this modification are displayed in Figure 4. Forming slots improved the AR values to satisfy down to the 1.4 GHz band from previously 1.5 GHz. Figure 4a,b shows the changes in S₁₁ and AR resulting from varying the length of the slots, confirming that the AR values at 1.4 GHz improve. However, manipulating the slots larger deteriorate the AR in the 2–2.5 GHz band. This issue can be resolved by rotating the crossed dipole. Figure 4c,d shows the results of S₁₁ and AR changes when the crossed dipole's rotation angle is varied, with the sidewall slot length fixed at 4 mm. Rotating the crossed dipole increases the AR above 2.9 GHz but relieves it in the lower bands. The crossed dipole was rotated by 15° in anticlockwise direction to generate stable coupling between the sidewall and the crossed dipole and ultimately, a single antenna operating from 1.4 to 3.7 GHz was designed. This process was carried out to have a more stable AR within the operating band for robustness.

FIGURE 4

The crossed dipole parameter study result: (a) S₁₁ regarding the side wall slot length. (b) Axial ratio regarding the side wall slot length. (c) S₁₁ regarding the crossed dipole rotation. (d) Axial ratio regarding the crossed dipole rotation.

The proposed crossed dipole antenna operates primarily through the crossed dipole part at the substrate and the ground reflector part formed below it. To clearly demonstrate this operation, the current distribution at t = 0 and t = T/4 for the proposed crossed dipole antenna was shown at Figure 5. The illustration shows the current distribution in the crossed dipole and the ground, which shifts clockwise as time varies. The current distribution on the ground is oriented from E-field coupling between the crossed dipole and the side metallic wall. This feature can be seen in Figure 6a, which represent the E-field distribution of xyz-plane on a height of 22 mm at t = 0 and t = T/4, respectively. At t = 0, the E-field couples with the side wall along the x-axis, and at t = T/4, it couples with the side wall along the y-axis. As the frequency decreases, the coupling becomes clear, so the E-field distribution at 1.4 GHz is shown to clarify the results.

FIGURE 5

Current distribution: (a) 1.4 GHz. (b) 2.4 GHz. (c) 3.6 GHz.

FIGURE 6

E-field distribution at 1.4 GHz: (a) Final design's E-field distribution in the xyz-plane at h = 22 mm. (b) E-field distribution of the design without a slot in the xyz-plane at h = 10 mm. (c) E-field distribution of the design with a slot in the xyz-plane at h = 10 mm.

However, the E-field that passes through the side wall creates an asymmetric field distribution at 1.4 GHz, which is lowest frequency of the operating band. A solution to this problem is to create slots in the side wall. Figure 6b,c show the xyz-plane field distribution at a height of 10 mm, illustrating the results without and with slots, respectively. The results show that in the design without slots, an asymmetric field distribution forms, but that with slots, a symmetric field distribution is achieved depending on the phase difference.

The effect of rotating the crossed dipole could be understood through the current distribution on the ground shown in Figure 5. Fundamentally, the current distribution along the direction of y = −x on the ground is almost negligible in the high-frequency region and strengthens as it shifts to lower frequencies, forming a current distribution of almost the same magnitude along the y = x direction. Additionally, the current distribution on the ground forms along the boundary of the crossed dipole, and rotating the crossed dipole makes each boundary trapezoidal. This structurally compensates for the asymmetry in the existing current distribution, thereby mitigating the AR in the mid-frequency band. However, as the frequency increases, the electrical length changes, resulting in a different phase and exacerbating the asymmetry in the current distribution, which is assessed as a degradation in the AR. Forming slots and rotating the crossed dipole are processes to adjust the current distribution on the ground, and to achieve a stable AR across a broad bandwidth. It is necessary to manipulate the current distribution on the ground.

2.3 Single element results

To validate the designed antenna, a prototype was fabricated and measurements were carried out. Figure 7 shows an overview of the single-element antenna prototype and the measurement setup. Consistent with the design, the Crossed dipole was formed on a 51 mm RF-60A substrate with a thickness of 1.52 mm, and the reflector ground with side wall structure below it was fabricated as a single cavity structure. The Crossed dipole substrate and the cavity structure were combined using screws. The prototype was measured in a 7 m chamber. Comparison of the simulated and actual measurement results for the single antenna's S₁₁, AR, and Realised gain is in Figure 8. The S₁₁ values are maintained below −10 dB across the frequency range from 1.3 to 3.8 GHz. Both the prototype and the simulation identify resonance points near 1.5 GHz, 2.8 GHz, and 3.5 GHz for AR, indicating that the antenna operates as CP within the frequency range of 1.4–3.6 GHz. The maximum AR observed is less than 2.7 dB in the operating band. The Realised Gain of the antenna maintain a level above 3 dB throughout the entire operational bandwidth. Figure 9 displays the radiation pattern of the single antenna at 1.4 GHz, 2.4 GHz, and 3.6 GHz, comparing the Co-pol operation in LHCP pattern and Cross-pol operation in RHCP pattern with both simulation and measurement results. The gain difference between Co-pol and Cross-pol at the θ = 0° axis is over 15 dB, indicating good isolation feature.

FIGURE 7

Overview of the single element antenna prototype: (a) Perspective view of single element prototype. (b) Measurement setup of single element prototype.

FIGURE 8

1D Results of single element antenna: (a) S₁₁ results. (b) Realized Gain results. (c) Axial Ratio results.

FIGURE 9

Far-field results of single element: (a) ϕ = 0° cut at 1.4 GHz. (b) ϕ = 90° cut at 1.4 GHz. (c) ϕ = 0° cut at 2.4 GHz. (d) ϕ = 90° cut at 2.4 GHz. (e) ϕ = 0° cut at 3.6 GHz. (f) ϕ = 90° cut at 3.6 GHz.

3 ARRAY DESIGN AND BEAM SCANNING

In array configurations, mutual coupling between elements can degrade the AR performance compared to that of a single element. The sequential rotation technique is one method to achieve a stable AR. Various studies, including [28, 29], have demonstrated that its application in array antennas results in improved performance. Especially when a CP single elements are used for element of sequential rotation array, sequential rotation yields a higher array factor value, wider bandwidth, and enhanced steering performance. Therefore, to ensure stable performance, sequential rotation was applied during the array design.

3.1 Sequential rotation and sequential divider

To verify the array design with sequential rotation through actual measurements, a sequential power divider that generates phase differences of 90°, 180°, and 270° designed and fabricated. The designed Sequential divider can be seen in Figure 10. The Sequential divider consists of a divider part and a phase shifter part. The divider part utilised a Wilkinson divider structure. The Wilkinson divider is designed to operate over a 1:3 bandwidth ratio using 110 , 210, and 390 Ω resisters to ensure a stable bandwidth [30]. The phase shifter part adopted a Coupled microstrip structure to generate the phase differences [31]. The coupled microstrip line phase shifter achieves a 90° phase difference across a 1:3 bandwidth within the working band by exploiting the phase differences between ports in a micro-coupled strip structure and its corresponding microstrip line. The performance of the Sequential divider is represented in Figure 11. Errors in S₂₁ and phase at each port can adversely affect the performance of sequential rotation. However, an amplitude error of less than 4 dB and a phase error of less than 30° generally lead to only minor errors in configuring sequential rotation, as documented in ref. [29]. The results presented in Figure 11 reveal a maximum amplitude error of 1.2 dB and a maximum phase error of 20°, both of which fall within acceptable error boundaries.

FIGURE 10

Overview of the sequential divider.

FIGURE 11

The results of sequential divider: (a) S₁₁ result of sequential divider. (b) S₂₁ result of sequential divider. (c) phase differnece results of sequential divider.

3.2 4 × 4 array design

Although sequential rotation has secured a stable AR, further attenuation of the active reflection coefficient is necessary. To minimise such interference, several optimisations were implemented. The rotation angle of the crossed dipole was adjusted from the previously set 15° to −10°, and the length of the side walls was reduced from 26 to 15 mm. The side wall slot was excluded while reducing the length of the side wall. As the sequential rotation was applied to the 4 × 4 array, the entire array was divided into quadrants centred at the middle of the array, with each quadrant functioning as an individual sub-array. Detailed phase information and antenna shape are in Figure 12a,b which shows the shape before the transformation, Type 1, and the shape after the transformation, Type 2 respectively. Specifically, the element in the same sub-arrays within each quadrant were assigned the same phase, with a quadrature phase difference applied in a anticlockwise direction to adjacent sub-arrays to facilitate sequential rotation. The elements within the sub-arrays of the 4 × 4 array were installed in a rotated direction according to the phase applied to each sub-array, which is indicated by an arrow. As a result, each sub-array contains elements that are symmetrically aligned with those in the other sub-arrays.

FIGURE 12

4 × 4 array configuration with the phase information and Active S parameter result: (a) Type 1 configuration. (b) Type 2 configuration. (c) Active S-parameter results of type 1. (d) Active S-parameter results of type 2.

Performance improvements from configuration changes are evident in the Active S-parameter results. Since each sub-array is symmetrical to the origin and yields similar outcomes, the results for ports 1, 2, 5, and 6, which are in the 90-degree phase set, are presented. While the original configuration typically exhibited an Active S-parameter around −6 dB, the modified configuration achieved a peak of −6.3 dB at 1.4 GHz and maintained a level around −8 dB up to 3.8 GHz. Variation in Active S parameter values across different frequencies are illustrated in Figure 12c,d.

3.3 4 × 4 array results

The 4 × 4 array was constructed as a single structure. This structure consists of one ground reflector for the 4 × 4 array, onto which side metallic walls were attached to create a cavity structure. A single 204 mm substrate, etched with the shapes of each element, was then screwed onto the metallic walls to complete the 4 × 4 array structure. The completed 4 × 4 array prototype with the Sequential divider are shown in Figure 13 including the setup process for measuring the active S-parameter and full array pattern. MS46122B was used for the active S-parameter analysis. When measuring the full array pattern, the 4 × 4 array and sequential divider were connected using a 1 m cable. It is fixed on a two-tier jig and then mounted to the mast in the measurement chamber.

FIGURE 13

4 × 4 array configuration: (a) Measurement setup of 4 × 4 array for active S-parameter. (b) Measurement setup of 4 × 4 array for full array pattern.

Figure 14a represents the Active S-parameter results of a comparison of four elements in the top left corner of 4 × 4 array. The measured active S-parameter values are slightly higher than those observed in the simulation, but they still exhibit stable performance within the operational bandwidth of 1.4–3.6 GHz, with a maximum value of −6.2 dB. The measured AR and Realised gain results are compared with the simulation results and presented in Figure 14b and c. The solid line graph represents the simulation results, while the dashed line graph shows the measured results. It was confirmed that the AR was consistently below 1 dB level through the operating band, with a peak of 2.2 dB at 1.4 GHz. The realised gain exhibits a minimum value of 10.5 dBic at 1.4 GHz, and a maximum value of 17.2 dBic at 3.6 GHz. The measurement results were adjusted for losses occurring at the Sequential divider. Figure 15 shows the radiation patterns of the 4 × 4 array at 1.4 GHz, 2.4 GHz, and 3.6 GHz. The radiation patterns are presented for both the xz-planes and yz-planes, ranging from −180° to 180°. The results exhibit similar tendencies to the simulation, with identical null points and gain values.

FIGURE 14

1D results of 4 × 4 array: (a) Active S-parameter results. (b) Axial ratio results. (c) Realised gain results.

FIGURE 15

Far-field results of 4 × 4 array: (a) ϕ = 0° cut at 1.4 GHz. (b) ϕ = 90° cut at 1.4 GHz. (c) ϕ = 0° cut at 2.4 GHz. (d) ϕ = 90° cut at 2.4 GHz. (e) ϕ = 0° cut at 3.6 GHz. (f) ϕ = 90° cut at 3.6 GHz.

3.4 Beam scanning results

Fundamentally, to electrically steer the beam in a phased array and prevent the occurrence of grating lobes, the distance between the phase centres of the elements must be kept below a certain threshold [25]. The condition for grating lobes can be observed in Equation (1) below. Here, λ₀ denotes the wavelength of the operating frequency, and θ₀ represents the beam steering direction. The element spacing d is determined by these two variables. The proposed antenna achieved wide grating lobe limitation area through miniaturisation. The Table 2 compares the designed element with other types of Crossed dipoles. The first size parameter (λ_l) provided in the table represents the overall size of the antenna based on the wavelength at the lowest operating frequency and the second size parameter (λ_h) corresponds to the wavelength at the highest operational frequency. From the table, it is noted that at low frequencies, the antennas dimensions are smaller than 0.5λ, which allows for steering up to 90°. However, at the high frequencies of the operational band, the beam scanning angle is significantly restricted due to the grating lobe. The crossed dipole referred to in ref. [21] is the smallest based on high-frequency standards, measuring 0.84λ. At this size, the maximum steering capability is limited to only 10°. The element referenced in ref. [20] operates over a broader bandwidth than the proposed antenna, and at 2.37 GHz, which matches the bandwidth of the proposed antenna, it can be steered up to 23° with a size of 0.71λ. This steering capability is better than that of the antenna in ref. [21]. However, the proposed antenna, with a size of 0.61λ at a high frequency, allows for maximum steering limitation of up to 39°, which is significantly greater than both refs. [20, 21]. A comparative analysis considering the actual beam steering performance of arrays formed using crossed dipole elements can be shown in Table 3. The proposed antenna is capable of beam steering and has a wide bandwidth of 88%.

TABLE 2. Comparison of crossed dipole.

Ref.	Size $$\left({\lambda }_{l}^{3}\right)$$	Size $$\left({\lambda }_{h}^{3}\right)$$	−10 dB IBW	3 dB ARBW	Technique
[20]	0.28 × 0.28 × 0.11	0.9 × 0.9 × 0.35	115.2% (0.84 ∼ 3.12 GHz)	106.1% (0.92 ∼ 3 GHz)	Coupled Rotated Vertical Metallic Plates
[21]	0.4 × 0.4 × 0.15	0.84 × 0.84 × 0.32	92.6% (1.15 ∼ 2.9 GHz)	71.8% (1.22 ∼ 2.65 GHz)	Shorted Vertical Plates
[22]	1.04 × 1.04 × 0.26	2.94 × 2.94 × 0.73	95.5% (0.92 ∼ 2.60 GHz)	94.4% (0.95 ∼ 2.65 GHz)	Cross-Dipole With Multiple Modes
[32]	0.46 × 0.46 × 0.10	0.89 × 0.89 × 0.19	78.3% (0.89 ∼ 2.03 GHz)	63.4% (0.97 ∼ 1.88 GHz)	Low-Profile Crossed-Dipole Antenna
[33]	0.79 × 0.79 × 0.27	1.44 × 1.44 × 0.49	68.9% (1.9 ∼ 3.9 GHz)	58.6% (2.05 ∼ 3.75 GHz)	Simple Single Parasitic Element
[34]	1.11 × 1.11 × 0.28	3.29 × 3.29 × 0.83	99.2% (1.24 ∼ 3.68 GHz)	72.7% (1.41 ∼ 3.02 GHz)	Parasitic Modified Patches
[35]	0.57 × 0.57 × 0.24	1.14 × 1.14 × 0.48	79.4% (1.9 ∼ 4.4 GHz)	66.7% (2 ∼ 4 GHz)	Dual-Cavity-Backed
Prop.	0.24 × 0.24 × 0.10	0.61 × 0.61 × 0.27	98.03% (1.3 ∼ 3.8 GHz)	88.0% (1.4 ∼ 3.6 GHz)	Koch-fractal crossed dipole

TABLE 3. Comparison of crossed dipole array.

Ref.	Element size $$\left({\lambda }_{h}^{3}\right)$$	Number of elements	Array size (mm^3)	Bandwidth	Electrical beam steering
[36]	0.97 × 0.97 × 0.26	2 × 2	200 × 200 × 27	38.2% (2 ∼ 2.91 GHz)	×
[37]	0.74 × 0.74 × 2.24	4 × 4	156 × 156 × 120	81.7% (2.35 ∼5.6 GHz)	×
[38]	1.02 × 1.02 × 0.47	2 × 2	140 × 140 × 23	104.4% (1.96 ∼6.24 GHz)	×
[39]	0.378 × 0.378 × 0.378	10 × 10	210 × 210 × 21	50% (2.7 ∼ 5.4 GHz)	○
Prop.	0.61 × 0.61 × 0.27	4 × 4	204 × 204 × 22	88% (1.4 ∼ 3.6 GHz)	○

Equation 1: Grating lobe condition

$$$d< \frac{{\lambda }_{0}}{1+\vert \sin \left({\theta }_{0}\right)\vert }$$$ (1)

To verify the beam scanning performance of the actual 4 × 4 array, measurement results of the active S-parameter were presented. [40] The active S-parameter results regarding the scanning angle are shown in Figure 16. The criterion for the active S parameter was set to −6 dB or less. [41] Measurements are provided for ± 30° in both the azimuth and elevation directions for ports 1, 2, 5, and 6, which are in the 90° phase set. Since the proposed antenna is symmetrical about the origin, the results for other ports, obtained through linear transformation, can be considered identical to those for ports 1, 2, 5, and 6. At 3 GHz, the value during 30° elevation steering reaches a maximum of −4.3 dB within the operating band; apart from this peak, the rest of the 12 ports exhibits values significantly below −6 dB.

FIGURE 16

Active S-parameter results of 4 × 4 array regarding scanning angle: (a) ϕ = 0° cut at port1. (b) ϕ = 90° cut at port1. (c) ϕ = 0° cut at port2. (d) ϕ = 90° cut at port2. (e) ϕ = 0° cut at port5. (f) ϕ = 90° cut at port5. (g) ϕ = 0° cut at port6. (h) ϕ = 90° cut at port6.

To observe the far-field patterns formed during beam steering, the Active Element Pattern (AEP) was measured, and the Array factor corresponding to the steering angle was multiplied by the measured AEP. The used AEP is essentially the average of all elements. Given that the proposed antenna is symmetrical about the origin, the AEP for the sub-array at a 90° phase set was measured separately, and the AEPs for the remaining elements were derived by linearly transforming the measured AEP by 90° in anticlockwise. The process of measuring the AEP is depicted in Figure 17, while Figure 18 displays the measured average AEP at 1.4, 2.4, and 3.6 GHz. Due to the symmetry of the antenna's shape about the origin, the patterns at ϕ = 0° and ϕ = 90° are identical. Figure 19 shows the beam scanning results after multiplying the measured average AEP by the array factor. The pattern displays the steering results at 15° and 30° degrees in both the azimuth and elevation directions, across frequencies of 1.4, 2.4, and 3.6 GHz. It is observed that as the beam scanning angle increases, the main lobe gain values decrease due to AEP losses. However, the positions of the lobes and the gain levels are formed similarly to the simulation results. These results demonstrate that the proposed array can achieve 30° steering in bandwidth of 1.4–3.6 GHz.

FIGURE 17

4 × 4 array configuration for measuring active element pattern: (a) Measurement setup of the 4 × 4 array configuration. (b) Overview of 4 × 4 array measurement for active element pattern.

FIGURE 18

Averaged AEP results: (a) 1.4 GHz. (b) 2.4 GHz. (c) 3.6 GHz.

FIGURE 19

Beam scanning results of 4 × 4 array: (a) ϕ = 0° cut at 1.4 GHz. (b) ϕ = 90° cut at 1.4 GHz. (c) ϕ = 0° cut at 2.4 GHz. (d) ϕ = 90° cut at 2.4 GHz. (e) ϕ = 0° cut at 3.6 GHz. (f) ϕ = 90° cut at 3.6 GHz.

4 CONCLUSIONS

The Koch folded fractal dipole is smaller than the conventional crossed dipole structure. The size of the single element antenna is 0.61λ × 0.61λ × 0.27λ at the higher frequency standard. The single element antenna proposed in this paper operates as a UWB CP antenna with a wide impedance matching band of about 98.03% from 1.3 to 3.8 GHz and an ARBW (AR Bandwidth) of 88% from 1.4 to 3.6 GHz. The single element was subsequently expanded, and using the sequential rotation technique, 4 × 4 array configurations were designed and validated. The validation of the 4 × 4 array configuration demonstrated that it is possible to miniaturise the antenna while suppressing mutual coupling, thereby enabling the array can steering up to 30°. The miniaturisation of the proposed antenna not only achieves a smaller size but also ensures beam scanning, with steering area of up to 30°. Including the phased array, the proposed miniaturised crossed dipole antenna could be applied to various CP UWB systems.

AUTHOR CONTRIBUTIONS

Won Min Choi: methodology; software; measurement; validation; formal analysis; writing—review and editing. Teayong Jeong: validation; data curation; draft preparation; writing—review and editing. Dong Hwan Lee: validation; validation. Jun Hee Kim: measurement. Dong Geun Lee: software. Keum Cheol Hwang: Conceptualisation; writing—review and editing; supervision; project administration; funding acquisition.

ACKNOWLEDGEMENTS

This work was supported by Korea Research Institute for defence Technology planning and advancement(KRIT) grant funded by the Korea government(DAPA(Defence Acquisition Program Administration)) (No. KRIT-CT-22-021, Space Signal Intelligence Research Laboratory, 2022).

CONFLICT OF INTEREST STATEMENT

The authors declare no potential conflicts of interest.

Open Research

DATA AVAILABILITY STATEMENT

The data that support the findings of this study are available from the corresponding author upon reasonable request.